The present invention relates generally to circuits and methods of compensating for variations in the current gain beta (β) of a transistor, and more particularly to circuits and methods of compensating for variations in the β of a single transistor used to measure temperature in an integrated circuit.
It is desirable to measure temperatures of silicon chips in various integrated circuit applications. One common method for achieving this is to measure a chip temperature using a single-temperature sensing transistor which has its base and emitter electrodes available for connection to temperature-measuring circuitry that may be on the same chip as the sensing transistor or on a different chip. The collector of the sensing transistor usually is connected to a reference voltage, such as ground or VDD.
“Prior Art” FIG. 1 shows a schematic diagram of the Assignee's commercially available TMP 411 remote temperature sensor integrated circuit product 1-1 which measures the junction temperature of a temperature-sensing vertical PNP transistor Q1 having its collector connected to ground. The emitter of sensing transistor Q1 is coupled by a conductor 2 and an anti-aliasing filter R5,R6,C2 to a (+) input of an integrating ADC (analog-to-digital converter) circuit 6 which produces a digital output on bus 7. The base of sensing transistor Q1 is coupled by conductor 3 and anti-aliasing filter R5,R6,C2 to a (−) input of ADC 6. A filter capacitor C1 is coupled between conductors 2 and 3. A current source I1 is coupled between VDD and conductor 2, and a current source I2 is coupled between VDD and conductor 3. A diode-connected PNP transistor Q2 has its emitter coupled to conductor 3 and its base and collector connected to ground. While this technique is simple and robust, it has the disadvantages that it is quite variable due to temperature variation of the chip and also is variable with the excitation current I1, especially for a low value of β.
In state-of-the-art integrated circuit manufacturing processes having line widths and spacings less than approximately 90 nanometers, the β (collector current divided by base current) of the sensing transistor (which typically is a vertical PNP “substrate” transistor) has a low, widely variable value. It is usually assumed that the β of sensing transistor Q1 (FIG. 1) is independent of its collector current. However, with line widths and spacings having values substantially less than 90 nanometers, that assumption becomes incorrect. The physics of a transistor shows that the base-emitter voltage VBE of the sensing transistor is basically a function of its collector current. If the current gain β of sensing transistor Q1 is not essentially independent of its collector current, rather than its emitter current, its base-emitter voltage VBE cannot be accurately calculated from the emitter current of that transistor.
In operation, current source 22 in FIG. 1 drives the current I2 through transistor Q2 in order to establish a bias voltage V3 on conductor 3. Current source I1 forces a first value of I1 into the emitter of sensing transistor Q1, causing the base current of sensing transistor Q1 to flow through conductor 3. The resulting collector current of sensing transistor Q1 flows into ground, causing a first value of VBE corresponding to the first value of I1. Integrating ADC 6 samples the first value of the base-emitter voltage VBE of sensing transistor Q1 after it has been filtered by anti-aliasing filter R5,R6,C2. Then a different second value of current I1 is forced through the emitter of sensing transistor Q1, which generates a second value of VBE voltage. Integrating ADC 6 samples that value of VBE and subtracts it from the first value of VBE. Thus, ADC 6 captures the difference between the two VBE voltages, and that VBE voltage difference is proportional to the absolute temperature of sensing transistor Q1.
The previous generations of “remote junction temperature sensor” integrated circuit products operate by controlling the emitter current, rather than the collector current, of the sensing transistor Q1, and use offset voltages and/or gain correction techniques to compensate for non-ideal characteristics of the sensing transistor. This prior technique of controlling the emitter current of the vertical PNP sensing transistor Q1 works well for systematic non-idealities, including the reduction of β and the increase in the variability of β as the spacings and line widths decrease substantially below 90 nanometers and also including various other non-ideal transistor parameter behavior, and has resulted in acceptably accurate temperature measurements. However, for present state-of-the-art manufacturing processes the chip-to-chip variation of the current gain β of the vertical PNP sensing transistor Q1 is great enough to cause large errors in the temperature measurements achieved by means of remote temperature sensor circuits such as the one shown in Prior Art FIG. 1. For example, present state-of-the-art manufacturing processes using line widths and spacings less than approximately 90 nanometers use doping levels that result in low, variable values of β which cannot be assumed to be independent of the collector current for the vertical PNP transistor Q1. Some prior art remote temperature sensor integrated circuits may utilize some kind of compensation for variations in the β of the temperature-sensing transistors thereof.
FIG. 2 shows the temperature measurement error in degrees Centigrade for a 1% change in β versus current gain β for the temperature measurement circuit 1-1 of Prior Art FIG. 1. Since sensing transistor Q1 typically is a vertical PNP with its collector connected to ground, it has been expedient to directly control the emitter current, as only the emitter and base electrodes of a vertical PNP transistor are accessible.
To avoid the effects of such increasing temperature measurement error caused by decreasing values of β in remote temperature sensor circuits such as the one shown in Prior Art FIG. 1 wherein the emitter current, rather than the collector current, of the vertical PNP sensing transistor Q1 is directly controlled, a circuit is needed which will control collector current instead of emitter current.
More specifically, the above mentioned variability of the β of the temperature-sensing transistor Q1 causes large errors in the temperature measurement values generated by the prior single-sensing-transistor temperature measurement circuits wherein precisely ratioed emitter currents are forced through the sensing transistor Q1 to generate a PTAT (proportional to absolute temperature) voltage ΔVBE, ΔVBE being the difference between the base-emitter VBE voltages of sensing transistor Q1 in response to the ratioed emitter currents. The large errors referred to are due to the fact that the base-emitter voltage VBE of a bipolar transistor depends directly on its collector current, and therefore current ratio values which are based on the collector current of the sensing transistor may be substantially different than corresponding current ratio values based on the emitter current thereof.
Thus, a major shortcoming of Prior Art FIG. 1 is that the emitter current of sensing transistor Q1, rather than the collector current, is what is directly controlled. As long as β is constant with respect to the controlled emitter current, or as long as β is fairly high in value, e.g., greater than approximately 10, the technique of Prior Art FIG. 1 provides a fairly accurate method of sensing the junction temperature of sensing transistor Q1, and the method of controlling the emitter current is acceptable. However, in modern integrated circuit manufacturing processes, the β of a vertical PNP transistor typically is very low, and furthermore, β changes as a function of current. So even though the ratio between a first emitter current and a second emitter current of sensing transistor Q1 is controlled, there is a completely different ratio between the corresponding first collector current and the corresponding second collector current. Therefore, the ratio of the first collector current to the second collector current is not accurately known. That causes significant errors in the ability of the circuit shown in Prior Art FIG. 1 to accurately measure the junction temperature of sensing transistor Q1, because only the emitter current, rather than the collector current, is directly controlled.
Thus, there is an unmet need for an improved circuit and method which use a single sensing transistor to measure the temperature of an integrated circuit chip wherein fabrication process characteristics cause low and variable values of the current gain β of the sensing transistor.
There also is an unmet need for a way of avoiding temperature measurement errors due to very low and/or widely varying values of the current gain β of a sensing transistor in an integrated circuit chip wherein fabrication process characteristics cause low and variable values of the current gain β of the sensing transistor.
There also is an unmet need for a way to precisely control the collector current in a temperature sensing transistor of a temperature sensor circuit wherein characteristics of the integrated circuit fabrication process being used cause low and variable values of the current gain β of the sensing transistor.
There also is an unmet need for a way of automatically estimating the current gain β of a temperature sensing transistor in an integrated circuit chip wherein fabrication process characteristics cause low and variable values of the current gain β of the sensing transistor.
There also is an unmet need for a way of automatically adjusting/optimizing various circuit parameters an integrated circuit chip wherein fabrication process characteristics cause low and variable values of the current gain β of a temperature sensing transistor.